Class D Audio Amplifier and Method for Reading a Current Supplied by the Amplifier

ABSTRACT

A circuit includes a final stage that includes an H-bridge comprising first and second half-bridges. A read circuit is configured to read a load current supplied by a class-D audio-amplifier to a load. The read circuit is configured for estimating the load current by reading a current at an output by the first or second half-bridge by measuring a drain-to-source voltage during an ON period of a power transistor of the H-bridge. A sensing circuit is configured to detect a first drain-to-source voltage from a transistor of the first half-bridge and a second drain-to-source voltage from a corresponding transistor of the second half-bridge. The sensing circuit is also configured to compute a difference between the first drain-to-source voltage and the second drain-to-source voltage and to perform an averaging operation on the difference to obtain a sense voltage value to be supplied to an analog-to-digital converter.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to Italian Application No. 102015000080985, filed on Dec. 7, 2015, which application is hereby incorporatedherein by reference.

TECHNICAL FIELD

Embodiments of the present invention relate to a class-D audio amplifiercomprising a circuit for reading a current supplied by the amplifier tothe load. Further embodiments relate to a corresponding reading method.

BACKGROUND

In class-D audio amplifiers, for various reasons, such as diagnostics ofthe state of speakers or else for applying thereto linearizationtechniques, there frequently arises the need to read accurately thecurrent that the final stage supplies to the load.

Since it is not generally convenient to resort to costly and cumbersomeexternal circuits for current sensing, recourse is had to internalsensing, by measuring the current supplied by power MOSFETs. At lowfrequency these currents are equal to the load current in so far as thecurrent that flows in filter capacitors is negligible.

In this connection, FIG. 1 illustrates a full-H bridge 11 of a finalstage of a power audio amplification circuit. The architecture of thefinal stage of a class-D amplification circuit is in itself known to theperson skilled in the sector, and in general comprises a comparator,which compares the input signal with a signal produced by atriangular-wave generator for supplying a PWM driving signal to aswitching controller, which in turn controls the states of opening andclosing of the MOSFETs of the H bridge 11, to the gate electrodes ofwhich it supplies the PWM driving signal. The H bridge 11 is a fullbridge, which comprises two half-bridges: a first half-bridge 12, whichcomprises a high-side power MOSFET 13 a, i.e., one connected to thesupply VDD, and a low-side power MOSFET 13 b, i.e., one connected toground GND, which supply, to a first output node OUTP that is commonbetween the drain electrode of the low-side transistor and the sourceelectrode of the high-side transistor, a first output current I_(OUT);and a second half-bridge 22, which comprises a respective high-sideMOSFET 23 a and a respective low-side MOSFET 23 b that supply to asecond output node OUTM a second current I_(OUTM).

The output current I_(OUTP), I_(OUTM) of each half-bridge 12, 22 of thebridge 11 is supplied, through a corresponding LC filter 14, 24, thefunction of which is also in itself known to the person skilled in thesector of class-D audio amplifiers and is not described any furtherherein, to the input terminals of a speaker 15, on which it determines aload current I_(LOAD). It should be noted that in FIG. 1 the high-sidetransistors are of an n-channel type, but they may also be of ap-channel type, in which case the output current is detected on thedrain electrode.

The solution currently adopted for carrying out reading of the loadcurrent I_(LOAD) is represented with reference to the circuit diagram ofFIG. 2 and to the corresponding time plot of FIG. 3.

It envisages sampling a drain-to-source voltage V_(DSLP) of the low-sidetransistor 13 b (or 23 b) of the circuit of FIG. 1 at the instant,designated by t_(s) in FIG. 3, in which the current ripple due to thefinite inductance of the external LC filter 14 (or 24) vanishes.

This is obtained via a sensing circuit 30, which comprises apre-amplifier 31 for carrying out reading of the drain-to-source voltageV_(DSLP) of the low-side transistor 13 b, the pre-amplifier 31 beingconnected with its two input pins to the drain and to the source of thelow-side transistor 13 b via driven protection switches 32 inserted oneach connection between the drain and source and the respective pins ofthe preamplifier 31. These switches 32 are driven by a protectionsignal, SWPRP, so as to protect the preamplifier 31 from high voltage,whilst further sampling switches set on the outputs of the preamplifier31 constitute, together with respective sampling capacitors, connectedto the analog ground AGND, a sample and hold circuit 33, driven by acorresponding sample and hold signal SWSH, which drives the states ofopening and closing of the switches of the circuit 33, for ananalog-to-digital converter (ADC) 34.

In this regard, FIG. 3 illustrates plots representing as a function oftime t the first output current I_(OUTP), the drain-to-source voltageV_(DSLP), the protection signal SWPRP, and the sample and hold drivingsignal SWSH. As may be noted, the drain-to-source voltage V_(DSLP) issampled at a sampling instant t_(s), where, at d.c. and at lowfrequencies, it is proportional to the load current I_(LOAD) through theswitch-on resistance R_(DSONLP) of the low-side transistor 13 b.

Moreover, designated by t_(a) in FIG. 3 is the instant of turning-off ofthe high-side MOSFET, for example 13 a, and turning-on of the low-sideMOSFET, for example 13 b. Designated by t_(b) is the instant ofturning-off of the low-side transistor and turning-on of the high-sidetransistor.

Once the drain-to-source voltage V_(DSLP) has been sampled, theinformation on current is obtained from the comparison with a referencedrain-to-source voltage V_(DSREF). This reference drain-to-sourcevoltage V_(DSREF) is generated by a reference power MOSFET 13 c(illustrated in FIG. 4), electrically and thermally coupled to theMOSFET 13 b that delivers the first output current I_(OUTP), with knownaspect ratio and current.

By computing the ratio between the drain-to-source voltage V_(DSLP) andthe reference drain-to-source voltage V_(DSREF), the first outputcurrent I_(OUTP) is obtained, as shown in FIG. 4, where designated byIREF is the current of a reference-current generator 16, which forcesthe current into the drain of the MOSFET 13 c. The drain-to-sourceswitch-on resistances the MOSFETs 13 b and 13 c are designated asR_(DSONLP) and R_(DSREF). The output current I_(OUTP) can thus becomputed according to these quantities as follows:

V _(DSREF) =R _(DSREF) ·I _(REF)

V _(DSLP) =R _(DSONLP)·(−I_(OUTP))

I _(OUTP)=−(I _(REF) ·R _(DSREF) /R _(DSONLP))·V _(DSLP) /V _(DSREF)

The drain-to-source voltages V_(DSLP) and V_(DSREF) are measured throughthe sensing circuit 30 of FIG. 2, whereas the other parameters are knowndesign parameters or are obtained following upon a further trimmingoperation.

The circuit just described with reference to FIGS. 2 and 3 presentsvarious limitations.

As shown in the plot of FIG. 5, which represents the output currentI_(OUTP) and a corresponding current I_(SAMP) sampled by the circuit 30,an error on the sampling instant t_(s) causes in fact a reading error.Ideally, sampling of the drain-to-source voltage V_(DSLP) must becarried out when the effect of the ripple in the inductance of the LCfilter 14 is zero and the output current I_(OUTP) is equal to the meanvalue of the load current I_(LOAD). If sampling is not made at thisinstant, but after a time Δt, superimposed on the signal is anundesirable current contribution ΔI due to the ripple current, asemerges from FIG. 5, represented in which are the current I_(LOAD) andtwo values of current I_(IDEAL) and I_(SAMP), one evaluated at the idealsampling time t_(IDEAL) and the other at the effective sampling timet_(SAMP)=t_(IDEAL)+Δt.

Sampling is moreover limited by saturation of the final stage.

When the signal grows, the duty cycle of the low-side transistordecreases (this time interval is represented by the high level of theprotection signal SWPRP in FIG. 3). When the duty cycle becomescomparable to or shorter than the time of charging of the gate, plus thesettling time both of the reading circuit and of the sample and holdcircuit, it is no longer possible to make an accurate reading of thecurrent. With reference to FIG. 3, this means that the time interval inwhich the sample and hold signal SWSH is high is too short. To preventthis problem linked to saturation, beyond a certain level of signal itwould be possible to choose to switch the reading of current on theother half-bridge, but switching of reading to the other half-bridgeentails the high risk of incurring in discontinuities.

SUMMARY

The present disclosure relates to a class-D audio amplifier comprising acircuit for reading a current supplied by the amplifier to a load, theamplifier including a final stage, which comprises an H bridge thatincludes a first half-bridge and a second half-bridge, the circuit forreading a load current being configured for estimating the load currentthrough reading of a current supplied at output by at least onehalf-bridge by measuring a drain-to-source voltage during an ON periodof at least one power transistor, in particular a low-side transistor,of the first half-bridge or the second half-bridge.

Various embodiments may be applied to audio power amplifiers, but alsoto other full-bridge stages that require detection of the load current.

Embodiments described herein improve the potential of the circuitsaccording to the known art, as discussed previously.

Various embodiments may envisage that the amplifier apparatus describedcomprises a sensing circuit including a circuit portion for detecting afirst drain-to-source voltage from a transistor of the firsthalf-bridge, and a second drain-to-source voltage from a correspondingtransistor of the second half-bridge. The sensing circuit includes amodule for computing a difference between the first drain-to-sourcevoltage and the second drain-to-source voltage that are detected by thecircuit portion, and also includes a module for performing an averagingoperation on the difference to obtain a sense voltage value to besupplied to an analog-to-digital converter.

Various embodiments may envisage that the module for computing thedifference between the first detected drain-to-source voltage and thesecond detected drain-to-source voltage is a differential amplifier.

Various embodiments may envisage that the module for performing anaveraging operation on the difference to obtain a sense voltage value isa low-pass filter.

Various embodiments may envisage that the circuit portion for detectinga first drain-to-source voltage from a transistor of the firsthalf-bridge and a second drain-to-source voltage from a correspondingtransistor of the second half-bridge comprises respective protectionswitches set between the drain and source nodes of the transistors ofthe first half-bridge and of the second half-bridge, and the respectiveinputs of the one module for computing the difference between the firstdetected drain-to-source voltage and the second detected drain-to-sourcevoltage.

Various embodiments may envisage that the circuit portion for detectinga first drain-to-source voltage from a transistor of the firsthalf-bridge and a second drain-to-source voltage from a correspondingtransistor of the second half-bridge comprises respective sensingnetworks, which include an auxiliary transistor, the gate and drain ofwhich are connected, respectively, to the gate and drain of therespective transistor of the first half-bridge or the secondhalf-bridge, and a sense resistance between the source of the auxiliarytransistor and the ground of the corresponding half-bridge.

Various embodiments may envisage that the sense resistance is sized sothat it has a value much higher than a switch-on resistance of theauxiliary transistor, such as to render the sum of the switch-onresistance of the auxiliary transistor and of the sense resistanceapproximately equal to the sense resistance.

Various embodiments may envisage that the sense resistance is sized sothat it has a value much lower than a switch-on resistance of theauxiliary transistor, such as to render the sum of the switch-onresistance of the auxiliary transistor and of the sense resistanceapproximately equal to the switch-on resistance.

Various embodiments may envisage that the apparatus comprises areference power MOSFET that generates a reference drain-to-sourcevoltage, which is electrically and thermally coupled to the powertransistor that delivers the output current, with known aspect ratio andcurrent.

Various embodiments may be aimed at providing a method for reading aload current supplied by a class-D audio amplifier to a load, whichcomprises detecting a first drain-to-source voltage from a transistor ofthe first half-bridge and a second drain-to-source voltage from acorresponding transistor of the second half-bridge, computing adifference between the first detected drain-to-source voltage and thesecond detected drain-to-source voltage, and performing an averagingoperation on the difference to obtain a sense voltage value to besupplied to an analog-to-digital converter.

BRIEF DESCRIPTION OF THE DRAWINGS

Various embodiments will now be described, purely by way of example,with reference to the annexed drawings, wherein:

FIG. 1 illustrates a full-H bridge 11 of a final stage of a power audioamplification circuit;

FIG. 2 illustrates a circuit diagram;

FIG. 3 illustrates a time plot corresponding to the circuit diagram ofFIG. 2;

FIG. 4 illustrates a reference power MOSFET;

FIG. 5 illustrates a plot that represents an output current and acorresponding sampled current;

FIG. 6 is a schematic illustration of a sensing circuit of the amplifierapparatus described;

FIGS. 7 and 8 show time plots of quantities in an amplifier apparatusthat uses the sensing circuit of FIG. 6;

FIG. 9 is a schematic illustration of a variant embodiment of thesensing circuit of FIG. 6;

FIG. 10 shows time plots of quantities in an amplifier apparatus thatuses the sensing circuit of FIG. 9;

FIG. 11 shows the sensing circuit of FIG. 9 according to a firstoperating configuration; and

FIG. 12 shows the sensing circuit of FIG. 9 according to a secondoperating configuration.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

In the ensuing description numerous specific details are provided inorder to enable maximum understanding of the embodiments provided by wayof example. The embodiments may be implemented with or without specificdetails, or else with other methods, components, materials, etc. Inother circumstances, well-known structures, materials, or operations arenot illustrated or described in detail so that various aspects of theembodiments will not be obscured. Reference, in the course of thisdescription, to “an embodiment” or “one embodiment” means that aparticular feature, structure, or characteristic described in connectionwith the embodiment is comprised in at least one embodiment. Hence,phrases such as “in an embodiment”, “in one embodiment”, or the likethat may be present in various points of the present description do notnecessarily refer to one and the same embodiment. Moreover, theparticular features, structures, or characteristics may be combined inany convenient way in one or more embodiments.

The notation and references are provided herein merely for convenienceof the reader and do not define the scope or meaning of the embodiments.

The idea underlying the solution described herein is to exploit theinformation of current of both of the half-bridges, and hence to detecta first drain-to-source voltage, preferably from a low-side transistorof the first half-bridge, and a second drain-to-source voltage from acorresponding low-side transistor of the second half-bridge. Thesampling operation is replaced by an operation of averaging on adifference between the first detected drain-to-source voltage and thesecond detected drain-to-source voltage, obtained, for example, via adifferential amplifier, to obtain a sense voltage value to be suppliedto an analog-to-digital converter. The averaging operation is preferablycarried out via a low-pass filtering of the aforethe difference betweenthe signal present on the MOSFET, for example the low-side MOSFET, ofthe H bridge during its ON period, and the signal present on thecorresponding transistor of the other half-bridge during its ON period.

Operation in the case of sensing performed on the low-side powertransistors is described with reference to FIGS. 6, 7, and 8.

FIG. 6 shows a block diagram, where a sensing circuit 50 comprises adifferential preamplifier 51, the input pins of which are connected,through switches 52P, a first one to the drain and a second one to thesource of the low-side MOSFET 13 b, and, through switches 52M, the firstone to the drain and the second one to the source of the low-side MOSFET23 b of the half-bridge 22. Downstream of the switches 52P and 52Mrespective first and second sense voltages V_(SP) and V_(SM) are formed,which correspond to the drain-to-source voltages detected, supplied tothe differential inputs of the preamplifier 51. The switches 52Pconnected to the first half-bridge 12 are driven by a protection signalSWPRP similar to that of FIG. 3, whereas the second switches 52M aredriven via a protection signal SVVPRM that has ON states, which arecomplementary in the case of out-of-phase modulation (FIG. 7), whereasin the case of in-phase modulation (FIG. 8) anyway coincide with the ONinterval of the respective low-side transistor (t_(aM) for the secondhalf-bridge). Hence, the protection switches 52P and 52M represent acircuit portion, or network, for detecting a first drain-to-sourcevoltage V_(SP) from a transistor 13 b of the first half-bridge 12 and asecond drain-to-source voltage V_(SM) from a corresponding transistor 23b of the second half-bridge 22 via respective protection switches 52P,52M set between the drain and source nodes of the transistors of thefirst half-bridge 12 and of the second half-bridge 22 and thepreamplifier 51. This preamplifier 51 provides a module for computing adifference between the first detected drain-to-source voltage V_(SP) andthe second detected drain-to-source voltage V_(SM).

Connected to the output of the preamplifier 51 is a low-pass filter 53,at the output of which a resulting sense voltage V_(SENSE) is formed.The low-pass filter 53 may also be integrated in the differentialamplifier 51 and hence not be a block cascaded thereto. The low-passfilter 53 provides a module for performing a continuous-time averagingof the aforethe difference to obtain a sense voltage value V_(SENSE) tobe supplied to an analog-to-digital converter 54.

In this regard, illustrated in FIGS. 7 and 8 are plots representing, asa function of time t, the first output current I_(OUTP), the secondoutput current I_(OUTM), the first sense voltage V_(SP), the secondsense voltage V_(SM), and the difference between the first sense voltageand the second sense voltage, V_(SM)-V_(SP). FIG. 7 shows the waveformsfor out-of-phase modulation, and FIG. 8 shows the waveforms for in-phasemodulation. As may be noted, the aforethe difference V_(sm)-V_(SP),which corresponds, through the amplification ratio, to the resultingsense voltage V_(SENSE), determines, through the low-pass filter 53, anaverage sense signal with value V_(S), both for the in-phase case andfor the out-of-phase case. Consequently, the sensing circuit 50 providesa continuous-time reading method that enables a greater accuracy in sofar as the averaging operation eliminates the contribution of the ripplewith zero mean value of the current in the finite inductance of the LCfilter 14 or 24.

It may be noted that it is moreover possible to read the current even inthe absence of the LC filter 14 or 24 in the circuit n of FIG. 1, asolution that eliminates the band limitations in so far as thismodification causes the load current I_(LOAD) to be equal to the firstoutput current I_(OUTP) and to the negative value of the second outputcurrent −I_(OUTM) not only at low frequencies.

Saturation of the reading system for high values of duty cycle isintrinsically eliminated.

Finally, it is possible to eliminate the circuits for generation of thesignals for the sample and hold circuit (SWSH in FIG. 2).

FIG. 9 illustrates a preferred alternative embodiment for generation ofthe first sense voltage V_(SP) and of the second sense voltage V_(SM) tobe supplied to the differential inputs of the preamplifier 51, whichenvisages replacement of each switch 52P and 52M with a respectivesensing network 62 shown in FIG. 9. This sensing network 62 comprises anauxiliary transistor 64, of the same type as the transistor 13 b or 23b, in general with an appropriate scale ratio that will guaranteethermal matching with the corresponding low-side transistor 13 b or 23b. The scale ratio is chosen as a compromise between occupation of areaand matching. The smaller the area ratio (or R_(DSON) ratio) between theMOSFET 13 b or 23 b and the auxiliary transistor 62, the better, ingeneral, the matching. With reference to what is referred to in whatfollows, this is valid when it is desired to operate in a partitioningmode, described hereinafter. In the case, instead, of operation in asensing mode, once again described in what follows, a good couplingbetween the MOSFET 13 b or 23 b and the auxiliary transistor 62 is nolonger necessary.

The gate and drain of the auxiliary transistor 64 are connected,respectively, to the gate and drain of the low-side transistor 13 b. Asense resistance R_(SP), sized for example as described hereinafter toobtain two different operating modes, is set between the source of theauxiliary transistor 64 and the ground GNDP of the half-bridge 12.Between the sensing resistances R_(SP) and R_(SM), respectively, of thetwo half-bridges and the preamplifier 51 a further amplification stagemay possibly be inserted.

Thanks to the added sensing network 62, during the OFF step of thelow-side transistor 13 b the sense signals V_(SP) and V_(SM) have avalue of o V instead of the value of the supply voltage VDD, asrepresented in FIG. 10, which shows the first output current I_(OUTP)and the first sense voltage V_(SP) as a function of time t, compatiblywith the operation described with reference to FIGS. 7 and 8.

As further advantage, the reading electronics is automatically protectedfrom the high voltage, and hence it is not necessary to resort tofurther protective circuitry, i.e., the switches 52P and 52M.

According to the mutual sizing between the switch-on resistanceR_(DSONAUXP) of the auxiliary transistor 64 and the sense resistanceR_(SP), it may possible to choose two different operating modes, takinginto account that for the circuit of FIG. 9 we have:

R_(DSONLP)<< R_(DSONAUXP) + R_(SP)V_(SP) = −I_(OUTP) ⋅ (R_(DSONLP) ⋅ R_(SP))/(R_(DSONLP) + R_(DSONAUXP) + R_(SP)) ≈ −I_(OUTP) ⋅ (R_(DSONLP) ⋅ R_(SP))/(R_(DSONAUXP) + R_(SP))

In a first, sensing, mode the sense resistance R_(SP) is sized so thatit has a value much higher than the switch-on resistance R_(DSONAUXP) ofthe auxiliary transistor 64, in particular such as to render the sum ofthe switch-on resistance R_(DSONAUXP) of the auxiliary transistor 64 andof the sense resistance R_(SP) approximately equal to the senseresistance R_(SP) in the relation appearing above that expresses thesense voltage V_(SP). In this way, the first sense voltage V_(SP) isapproximately equal to −I_(OUTP)·R_(DSONLP), and sensing of thedrain-to-source voltage V_(DSLP) of the low-side transistor 13 b iscarried out.

In a second, partitioning, mode, the sense resistance R_(SP) is sized sothat it has a value much lower than the switch-on resistanceR_(DSONAUXP), in particular such as to render the sum of the switch-onresistance R_(DSONAUXP) of the auxiliary transistor 64 and of the senseresistance R_(SP) approximately equal to the switch-on resistanceR_(DSONAUXP) in the relation appearing above that expresses the sensevoltage V_(SP). Hence, the first sense voltage V_(SP) is approximatelyequal to −I_(OUTP) 9 (R_(DSONLP)/R_(DSONAUXP))·R_(SP)), and thus thesensing operation is carried out on a partitioning of known value of thefirst output current I_(OUTP).

Of course, similar arguments apply in a dual way to the low-sidetransistor 23 b of the second half-bridge 22.

In greater detail, with reference to the diagram of FIG. 11, in thefirst, sensing, mode the drain-to-source voltage V_(DSLP) of thelow-side transistor 13 b is measured.

To obtain the information of current, as described with reference toFIG. 4, it is necessary to compare the measured drain-to-source voltageV_(DSLP) of the low-side transistor 13 b with the reference voltageV_(DSREF) resulting from a known current, the reference current I_(REF),that flows in the reference MOS transistor 13 c coupled to the low-sidetransistor 13 b according to the relation:

I_(OUT)≈−(I_(REF)·(R_(DSREF)/R_(DSONLP)))·(V_(SP)/V_(DSREF))

where (I_(REF)·(R_(DSREF)/R_(DSONLP))) is a term the values of which areknown design values and (V_(SP)/V_(DSREF)) is a term the values of whichare measured. More specifically, of the term (V_(SP)/V_(DSREF)) only thevalue of the ratio is measured, for example using an ADC, where thefull-scale is regulated by the reference drain-to-source voltageV_(DSREF), and the voltage to be converted is the sense voltage V_(SP),so that the output of the converter depends in actual fact only upon theratio between the two values.

The aforethe first, sensing, mode presents the following advantages:

no thermal matching between the low-side transistor 13 b (or 23 b) andthe auxiliary transistor 64 is required;

the layout is simplified;

the signal to be amplified is the maximum one available in so far as thedrain-to-source voltage V_(DSLP) of the low-side transistor is notpartitioned; and

the reading electronics is simplified.

Since the non-linearity of the drain-to-source voltage V_(DSLP) of thelow-side transistor as a function of current causes a non-linearity inthe reading, the first, sensing, mode is preferable in the cases ofrelatively low drain-to-source voltages, in which the effect ofnon-linearity is negligible.

With reference to the diagram of FIG. 12, in the second, partitioning,mode a partitioning of the current that flows in the low-side transistor13 b is measured.

To obtain the information of current, it is necessary to compare thesense voltage V_(SP) with a reference constituted by the known currentI_(REF) that flows in a reference resistance 17 of value R_(REF) coupledto the sense resistance R_(SP), i.e., connected between the generator 16of reference current I_(REF) and ground GNDP, on which there is areference voltage drop V_(REF), according to the relation:

I_(OUT)≈−(I_(REF)·(R_(REF)/R_(SP))·(R_(DSONAUXP)/R_(DSONLP)))·(V_(SP)/V_(REF)),

where (I_(REF)·(R_(REF)/R_(SP))·(R_(DSONAUXP)/R_(DSONLP))) is a term thevalues of which are known design values, and (V_(SP)/V_(REF)) is a termthe values of which, or the value of their ratio, are/is measured. Itshould be noted that to obtain a measurement of I_(OUT) independent oftemperature and process, the measurement must depend upon ratios ofresistances of the same type, so that the process or temperaturevariations cancel out, the measurements amounting only to ratios ofareas. Hence, R_(REF)/R_(SP) is a ratio between values of two resistors,R_(DSONAUXP)/R_(DSONLP) is a ratio between values of two MOSFETs in theohmic region (the sense resistance R_(SP) must be much lower than theswitch-on resistance R_(DSONLP) of the low-side transistor 13 b so thatthe power MOSFET 13 b and the auxiliary MOS transistor 62 work as far aspossible in the same condition, i.e., the same gate-to-source voltageV_(Gs) and the same drain-to-source voltage V_(SP)).

This second, partitioning, mode presents, as compared to the first mode,the advantage that the effect of the non-linearity of thedrain-to-source voltage of the low-side transistor is limited, and hencethe measurement can be very linear.

On the other hand, accurate thermal and electrical matching is requiredbetween the low-side transistor, for example 13 b, and the auxiliarytransistor 64, so as to guarantee a well-controlled partitioning. Forthis reason, the layout of the power transistors is more complex. Thereis also required good thermal and electrical matching between the senseresistors R_(SP) and R_(REF). The sense resistance R_(SP) may be of avery small value so as to be negligible as compared to the resistance ofthe auxiliary switch, this possibly resulting in a complex layout of thesense resistance, whilst the signal to be amplified may be very small,thus rendering the reading electronics more critical.

The second, partitioning, mode is hence preferable in the cases wherethe effect of the non-linearity of the drain-to-source voltage is notacceptable. Instead, the first, sensing, mode in any case presentsadvantages in terms of complexity of the layout and simplicity of thereading circuit.

Hence, the advantages of the solution described emerge clearly from theforegoing description.

The class-D audio amplifier comprising a circuit for reading a currentsupplied by the amplifier to the load described herein advantageouslyprovides a continuous-time reading method that enables a greateraccuracy in so far as the operation of averaging eliminates thecontribution of the zero-mean ripple of the current in the finiteinductance of the LC filter.

Moreover, this amplifier advantageously enables reading of the currenteven in the absence of the LC filter, thus eliminating the bandlimitations.

Furthermore, this amplifier advantageously enables intrinsic eliminationof the saturation of the reading system for high values of duty cycle.

Finally, this amplifier advantageously enables elimination of thecircuits for generation of signals for the sample and hold circuit.

In addition, the use of a sensing network in the amplifier describedenables automatic protection of the reading electronics from highvoltage.

Moreover, advantageously, via simple sizing of a sense resistance, thesensing network is readily configurable for use in sensing mode withsimpler layout and reading electronics, or in partitioning mode, whichis less sensitive to non-linear behaviours. In particular, dependence ofthe measurement upon the non-linearity of the drain-to-sourcecurrent-voltage characteristic of the MOSFET of the half-bridge,acquired on which is the output current, is eliminated.

Of course, without prejudice to the principle of the invention, thedetails and the embodiments may vary, even considerably, with respect towhat has been described herein purely by way of example, without therebydeparting from the sphere of protection, which is defined in the annexedclaims.

The class-D audio amplifier apparatus comprising a circuit for reading aload current supplied by the amplifier apparatus to a load describedherein may envisage reading, for estimating the load current, a currentsupplied at output by a half-bridge by measuring a drain-to-sourcevoltage (V_(DSLP)) during an ON period of the high-side (n-channel orp-channel) power MOSFETs, instead of carrying out the measurement on thelow-side ones, even though it is in general more convenient to operatewith a circuit referenced to ground, rather than with a circuitreferenced to the supply voltage. In the case of high-side p-channeltransistors a complementary equivalent solution is used, with thesources connected to the supply voltage.

What is claimed is:
 1. A circuit comprising a final stage that includesan H bridge comprising a first half-bridge and a second half-bridge; aread circuit configured to read a load current supplied by a class-Daudio-amplifier to a load, the read circuit being configured forestimating the load current by reading a current at an output by thefirst or second half-bridge by measuring a drain-to-source voltageduring an ON period of a power transistor of the first half-bridge orsecond half-bridge; and a sensing circuit comprising a circuit portionconfigured to detect a first drain-to-source voltage from a transistorof the first half-bridge and a second drain-to-source voltage from acorresponding transistor of the second half-bridge, the sensing circuitcomprising a first sub-circuit configured to compute a differencebetween the first drain-to-source voltage and the second drain-to-sourcevoltage, which are detected by the circuit portion, and a secondsub-circuit configured to perform an averaging operation on thedifference to obtain a sense voltage value to be supplied to ananalog-to-digital converter.
 2. The circuit according to claim 1,wherein the power transistor comprises a low-side transistor of thefirst half-bridge or the second half-bridge.
 3. The circuit according toclaim 1, wherein the first sub-circuit comprises a differentialamplifier.
 4. The circuit according to claim 1, wherein the secondsub-circuit comprises a low-pass filter.
 5. The circuit according toclaim 1, wherein the circuit portion comprises respective protectionswitches set between drain and source nodes of the transistors of thefirst half-bridge and of the second half-bridge and inputs of the firstsub-circuit.
 6. The circuit according to claim 1, wherein the circuitportion comprises a sensing network includes an auxiliary transistorwith a gate and a drain, wherein the gate of the auxiliary transistor iscoupled to the gate of a respective transistor of the first half-bridgeor second half-bridge and wherein a drain of the auxiliary transistor iscoupled to the drain of the respective transistor of the firsthalf-bridge or second half-bridge.
 7. The circuit according to claim 6,wherein the circuit portion further comprises a sense resistance betweena source of the auxiliary transistor and ground of a correspondinghalf-bridge.
 8. The circuit according to claim 7, wherein the senseresistance is sized so that the sense resistance has a higher value thana switch-on resistance of the auxiliary transistor, such as to render asum of the switch-on resistance of the auxiliary transistor and of thesense resistance approximately equal to the sense resistance.
 9. Thecircuit according to claim 7, wherein the sense resistance is sized sothat the sense resistance has a lower value than a switch-on resistanceof the auxiliary transistor, such as to render a sum of a switch-onresistance of the auxiliary transistor and of the sense resistanceapproximately equal to the switch-on resistance.
 10. The circuitaccording to claim 7, further comprising a reference power MOSFETconfigured to generate a reference drain-to-source voltage, thereference power MOSFET being electrically and thermally coupled to thepower transistor that supplies the current at the output with a knownaspect ratio and current.
 11. A method for reading a load currentsupplied by the class-D audio-amplifier to the load using the circuitaccording to claim
 1. 12. A method for reading a load current suppliedby a class-D audio amplifier to a load by estimating the load current,the method comprising: reading a current supplied by a power transistorof a first half-bridge or a second half-bridge of an H bridge of a finalstage of the amplifier; detecting a first drain-to-source voltage from atransistor of the first half-bridge; detecting a second drain-to-sourcevoltage from a corresponding transistor of the second half-bridge;computing a difference between the first detected drain-to-sourcevoltage and the second detected drain-to-source voltage; and performingan averaging operation on the difference to obtain a sense voltagevalue.
 13. The method according to claim 12, further comprisingobtaining the load current by comparing the sense voltage value with areference drain-to-source voltage.
 14. The method according to claim 12,further comprising obtaining the load current by comparing the sensevoltage value with a voltage on a reference resistor.
 15. The methodaccording to claim 12, wherein reading the current comprises reading thecurrent supplied by a low-side transistor of the first half-bridge orthe second half-bridge of the H bridge of a final stage of theamplifier.
 16. The method according to claim 12, wherein reading thecurrent comprises reading a current supplied by a power transistor ofthe first half-bridge and also reading a current supplied by a powertransistor of the second half-bridge.
 17. The method according to claim12, further comprising supplying the sense voltage value to ananalog-to-digital converter.
 18. A circuit comprising a final stage thatincludes an H bridge comprising a first half-bridge and a secondhalf-bridge; a read circuit configured to read a load current suppliedby a class-D audio-amplifier to a load, the read circuit beingconfigured to estimate the load current by reading a current at anoutput by the first or second half-bridge by measuring a drain-to-sourcevoltage during an ON period of a low-side transistor of the firsthalf-bridge or second half-bridge; and a sensing circuit comprising adifferential amplifier and a low-pass filter, the sensing circuitconfigured to detect a first drain-to-source voltage from a transistorof the first half-bridge and a second drain-to-source voltage from acorresponding transistor of the second half-bridge, wherein thedifferential amplifier is configured to compute a difference between thefirst drain-to-source voltage and the second drain-to-source voltage,wherein the low-pass filter is configured to perform an averagingoperation on the difference to obtain a sense voltage value.
 19. Thecircuit according to claim 18, wherein sensing circuit comprisesrespective protection switches set between drain and source nodes of thetransistors of the first half-bridge and of the second half-bridge andinputs of the differential amplifier.
 20. The circuit according to claim18, wherein the sensing circuit comprises a sensing network thatincludes an auxiliary transistor with a gate and a drain, wherein thegate of the auxiliary transistor is coupled to a gate of a respectivetransistor of the first half-bridge or second half-bridge and wherein adrain of the auxiliary transistor is coupled to a drain of therespective transistor of the first half-bridge or second half-bridge.21. The circuit according to claim 20, wherein the sensing circuitfurther comprises a sense resistance between a source of the auxiliarytransistor and ground of a corresponding half-bridge.
 22. The circuitaccording to claim 21, wherein the sense resistance is sized so that thesense resistance has a higher value than a switch-on resistance of theauxiliary transistor, such as to render a sum of the switch-onresistance of the auxiliary transistor and of the sense resistanceapproximately equal to the sense resistance.
 23. The circuit accordingto claim 22, wherein the sense resistance is sized so that the senseresistance has a lower value than a switch-on resistance of theauxiliary transistor, such as to render a sum of a switch-on resistanceof the auxiliary transistor and of the sense resistance approximatelyequal to the switch-on resistance.